Low power loss current digital-to-analog converter used in an implantable pulse generator

ABSTRACT

In one embodiment, the present invention provides an implantable stimulation device that includes output current sources and/or sinks configured to provide an output current for a load (i.e., tissue). The output path of the output current source or sink comprises a transistor which operates in a linear mode instead of a saturation mode. Because operation in a linear mode results in smaller drain-to-source voltage drops, power consumption in the output current source or sink (and hence in the implantable stimulator) is reduced, reducing battery or other power source requirements. Operation in the linear mode is facilitated in useful embodiments by a load in an input path (into which a reference current is sent) and a load in the output path (which bears the output current). The loads can be active transistors or passive resistors. A feedback circuit (e.g., an operational amplifier) receives voltages that build up across these loads, and sends a control signal to the gate of the transistor to ensure its linear operation.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a non-provisional application claiming prioritypursuant to 35 U.S.C. § 119(e) to U.S. Provisional Patent ApplicationSer. No. 60/575,725, filed May 28, 2004, which is incorporated herein byreference in its entirety.

FIELD OF THE INVENTION

The present invention relates generally to implantable pulse generators,e.g., a pulse generator used within a Spinal Cord Stimulation (SCS)system or other type of neural stimulation system. More particularly,the present invention relates to the use of output current sourceshaving a current digital to analog converter (DAC) configured toregulate the current delivered via an implantable pulse generator (IPG).

BACKGROUND

Implantable stimulation devices are devices that generate and deliverelectrical stimuli to body nerves and tissues for the therapy of variousbiological disorders, such as pacemakers to treat cardiac arrhythmia,defibrillators to treat cardiac fibrillation, cochlear stimulators totreat deafness, retinal stimulators to treat blindness, musclestimulators to produce coordinated limb movement, spinal cordstimulators to treat chronic pain, cortical and deep brain stimulatorsto treat motor and psychological disorders, and other neural stimulatorsto treat urinary incontinence, sleep apnea, shoulder sublaxation, etc.The present invention may find applicability in all such applications,although the description that follows will generally focus on the use ofthe invention within a spinal cord stimulation system, such as thatdisclosed in U.S. Pat. No. 6,516,227, issued Feb. 4, 2003 in the name ofinventors Paul Meadows et al., which is incorporated herein by referencein its entirety.

Spinal cord stimulation is a well accepted clinical method for reducingpain in certain populations of patients. A Spinal Cord Stimulation (SCS)system typically includes an Implantable Pulse Generator (IPG) orRadio-Frequency (RF) transmitter and receiver, electrodes, at least oneelectrode lead, and, optionally, at least one electrode lead extension.The electrodes, which reside on a distal end of the electrode lead, aretypically implanted along the dura of the spinal cord, and the IPG or RFtransmitter generates electrical pulses that are delivered through theelectrodes to the nerve fibers within the spinal column. Individualelectrode contacts (the “electrodes”) are arranged in a desired patternand spacing to create an electrode array. Individual wires within one ormore electrode leads connect with each electrode in the array. Theelectrode lead(s) exit the spinal column and generally attach to one ormore electrode lead extensions. The electrode lead extensions, in turn,are typically tunneled around the torso of the patient to a subcutaneouspocket where the IPG or RF receiver is implanted. Alternatively, theelectrode lead may directly connect with the IPG or RF receiver.

SCS and other stimulation systems are known in the art. For example, animplantable electronic stimulator is disclosed in U.S. Pat. No.3,646,940, issued Mar. 7, 1972, entitled “Implantable ElectronicStimulator Electrode and Method,” which teaches timed sequencedelectrical pulses to a plurality of electrodes. Another example, U.S.Pat. No. 3,724,467, issued Apr. 3, 1973, entitled “Electrode Implant forthe Neuro-Stimulation of the Spinal Cord,” teaches an electrode implantfor neuro-stimulation of the spinal cord. A relatively thin and flexiblestrip of biocompatible material is provided as a carrier on which aplurality of electrodes reside. The electrodes are connected by aconductor, e.g., a lead body, to an RF receiver, which is also implantedand is controlled by an external controller.

U.S. Pat. No. 3,822,708, issued Sep. 9, 1974, entitled “ElectricalSpinal Cord Stimulating Device and Method for Management of Pain,”teaches an SCS device with five aligned electrodes which are positionedlongitudinally along the spinal cord. Current pulses applied to theelectrodes block sensed intractable pain, while allowing passage ofother sensations. The stimulation pulses applied to the electrodes havea repetition rate of 5 to 200 pulses per second. A patient-operatedswitch allows the patient to change the electrodes that are activated(i.e., which electrodes receive the stimulation pulses) to stimulate aspecific area of the spinal cord, as required, to better block the pain.

Regardless of the application, all implantable pulse generators areactive devices requiring energy for operation. The energy is supplied bya power source that may be an implanted battery or an external powersource. It is desirable for the implantable pulse generator to operatefor extended periods of time with little intervention by the patient orcaregiver. However, devices powered by primary (non-rechargeable)batteries have a finite lifetime before the device must be surgicallyremoved and replaced. Frequent surgical replacement is not an acceptablealternative for many patients. If a battery is used as the energysource, it must have a large enough storage capacity to operate thedevice for a reasonable length of time. For low-power devices (less than100 μW) such as cardiac pacemakers, a primary battery may operate for areasonable length of time, often up to ten years. However, in manyneural stimulation applications such as SCS, the power requirements areconsiderably greater due to higher stimulation rates, pulse widths, orstimulation thresholds.

Thus, one challenge with IPGs is keeping power usage to a minimum toconserve battery life. While increasing battery life may be achieved byextending the size of the battery, that runs counter to the goal ofreducing the overall device size which is determined partly by batterysize. Conservation of energy in an implantable, battery operated deviceis an important design goal to reduce the overall size of the device andto prolong the life of the battery, thus deferring surgery to replacethe device.

An IPG often includes one or more output current sources that areconfigured to supply current to a load, such as tissue, associated withthe IPG. The output current source may include a current digital toanalog converter (DAC) configured to regulate the current that isdelivered to the load. However, the DAC is often physically located inseries with the load. Hence, any load current passes through the DAC aswell, which results in a power loss. This power loss may result in ashortening of the battery life of the IPG. The power loss is directlyproportional to the voltage drop across the DAC. Accordingly, there is agreat need for an IPG having an output current source that includes acurrent DAC having a small voltage drop such that the power efficiencyof the IPG is maximized.

SUMMARY

In one embodiment, the present invention provides an implantablestimulation device that includes output current sources and/or sinksconfigured to provide an output current for a load. The output path ofthe output current source or sink comprises a transistor which operatesin a linear mode instead of a saturation mode. Because operation in alinear mode results in smaller drain-to-source voltage drops, powerconsumption in the output current source or sink (and hence in theimplantable stimulator) is reduced, reducing battery or other powersource requirements. Operation in the linear mode is facilitated inuseful embodiments by a load in an input path (into which a referencecurrent is sent) and a load in the output path (which bears the outputcurrent). The loads can be active transistors or passive resistors. Afeedback circuit (e.g., an operational amplifier) receives voltages thatbuild up across these loads, and sends a control signal to the gate ofthe transistor to ensure its linear operation.

In preferred embodiments, the output current sources or sinks comprise adigital-to-analog converter (DAC) for scaling the magnitude of theoutput current versus the input reference current. Moreover, in otherembodiments, the values of the loads are varied (by a ratio of N) toallow the output current to be amplified with respect to the referencecurrent, without the necessity of adding further output stages inparallel.

BRIEF DESCRIPTION OF THE DRAWINGS

The above and other aspects of the present invention will be moreapparent from the following more particular description thereof,presented in conjunction with the following drawings wherein:

FIG. 1 shows a block diagram that illustrates the various implantable,external, and surgical components of a spinal cord stimulation (SCS)system that employs an implantable pulse generator (IPG) having arechargeable battery in accordance with the present invention;

FIG. 2 shows various components of the SCS system of FIG. 1;

FIG. 3 shows a block diagram that illustrates the main components,including a rechargeable battery, of one embodiment of an implantablepulse generator (IPG) used with the invention;

FIG. 4 shows a block diagram that illustrates another embodiment of animplantable pulse generator (IPG) that may be used with the invention;

FIG. 5 shows an exemplary output current source and a correspondingoutput current sink each having current digital-to-analog converter(DAC) circuitry in series with a load;

FIG. 6 shows a modified design for the output current source and sink ofFIG. 5 in accordance with an embodiment of the present invention;

FIGS. 7A and 7B show how use of the design of the output current sourceor sink of FIG. 6 (FIG. 7B) minimizes layout complexity as compared tothe output current source or sink of FIG. 5 (FIG. 7A);

FIG. 8 shows another modified design for the output current source andsink of FIG. 5 in accordance with an embodiment of the presentinvention; and

FIG. 9 shows the design of the improved output current source or sinkillustrated generically with loads and a feedback circuit.

Corresponding reference characters indicate corresponding componentsthroughout the several views of the drawings.

DETAILED DESCRIPTION

The following description is of the best mode presently contemplated forcarrying out the invention. This description is not to be taken in alimiting sense, but is made merely for the purpose of describing thegeneral principles of the invention. The scope of the invention shouldbe determined with reference to the claims.

At the outset, it is noted that the present invention may be used withan implantable pulse generator (IPG), or similar electrical stimulatorand/or electrical sensor, that may be used as a component of numerousdifferent types of stimulation systems. The description that followsrelates to use of the invention within a spinal cord stimulation (SCS)system. However, it is to be understood that the invention is not solimited. Rather, the invention may be used with any type of implantableelectrical circuitry that could benefit from an output current sourcehaving a current DAC topology configured to maximize energy efficiency.For example, the present invention may be used as part of a pacemaker, adefibrillator, a cochlear stimulator, a retinal stimulator, a stimulatorconfigured to produce coordinated limb movement, a cortical and deepbrain stimulator, or in any other neural stimulator configured to treaturinary incontinence, sleep apnea, shoulder sublaxation, etc.

Turning first to FIG. 1, a block diagram is shown that illustrates thevarious components of an exemplary SCS system in which the invention maybe used. These components may be subdivided into three broad categories:(1) implantable components 10, (2) external components 20, and (3)surgical components 30. As seen in FIG. 1, the implantable components 10include an implantable pulse generator (IPG) 100, an electrode array110, and (as needed) a lead extension 120. The extension 120 may be usedto electrically connect the electrode array 110 to the IPG 100. In anexemplary embodiment, the IPG 100, described more fully below inconnection with FIG. 3 or 4, may comprise a rechargeable, multichannel,sixteen contact, telemetry-controlled, pulse generator housed in arounded high-resistivity titanium alloy case to reduce eddy currentheating during the inductive charging process. The embodiment mayinclude sixteen current sources, each with a programmable amplitude suchthat the device is a current-regulated, rather than a voltage-regulatedsystem. However, it will be noted that in an alternative embodiment, aSCS system may include more or less than sixteen current sources.

According to an exemplary embodiment of the present invention, the IPG100 may include stimulating electrical circuitry (“stimulatingelectronics”), a power source, e.g., a rechargeable battery, and atelemetry system. Typically, the IPG 100 is placed in a surgically-madepocket either in the abdomen, or just at the top of the buttocks. Itmay, of course, also be implanted in other locations of the patient'sbody. Once implanted, the IPG 100 is connected to the lead system,comprising the lead extension 120, if needed, and the electrode array110. The lead extension 120, for example, may be tunneled up to thespinal column. Once implanted and any trial stimulation period iscomplete, the lead system 110 and lead extension 120 are intended to bepermanent. In contrast, the IPG 100 may be replaced when its powersource fails or is no longer rechargeable.

Advantageously, the IPG 100 may provide electrical stimulation through amultiplicity of electrodes, e.g., sixteen electrodes, included withinthe electrode array 110.

As seen best in FIG. 2, and as also illustrated in FIG. 1, the electrodearray 110 and its associated lead system typically interface with theimplantable pulse generator (IPG) 100 via a lead extension system 120.The electrode array 110 may also be connected to an external trialstimulator 140, through the use of a percutaneous lead extension 132and/or an external cable 134. The external trial stimulator 140 includesthe same pulse generation circuitry as does the IPG 100, and is used ona trial basis for, e.g., 7-10 days after the electrode array has beenimplanted, prior to implantation of the IPG 100, to test theeffectiveness of the stimulation that is to be provided.

Still with reference to FIGS. 1 and 2, the hand-held programmer (HHP)202 may be used to control the IPG 100 via a suitable non-invasivecommunications link 201, e.g., an RF link. Such control allows the IPG100 to be turned ON or OFF, and generally allows stimulation parameters,e.g., pulse amplitude, width, and rate, to be set within prescribedlimits. The HHP 202 may also be linked with the external trialstimulator 140 through another link 205′, e.g., an infra red link.Detailed programming of the IPG 100 is preferably accomplished throughthe use of an external clinician's programmer 204 (FIG. 1), which mayalso be hand-held and which may be coupled to the IPG 100 directly orthrough the HHP 202. An external charger 208, non-invasively coupledwith the IPG 100 through link 209, e.g., an inductive link, allowsenergy stored or otherwise made available to the charger 208 to becoupled into the rechargeable battery housed within the IPG 100.

Turning next to FIG. 3, a block diagram is shown that illustrates themain components of one embodiment of an implantable pulse generator(IPG) 100 that may be used with the invention. As seen in FIG. 3, theIPG may include a microcontroller (μC) 160 connected to memory circuitry162. The μC 160 typically comprises a microprocessor and associatedlogic circuitry, which in combination with control logic circuits 166,timer logic 168, and an oscillator and clock circuit 164, generate thenecessary control and status signals which allow the μC 160 to controlthe operation of the IPG in accordance with a selected operating programand stimulation parameters. The operating program and stimulationparameters are typically stored within the memory 162 by transmitting anappropriate modulated carrier signal through a receiving coil 170 andcharging and forward telemetry circuitry 172 from an externalprogramming unit, e.g., a handheld programmer 202 and/or a clinicianprogrammer 204, assisted as required through the use of a directionaldevice 206 (see FIG. 1). (The handheld programmer is thus considered tobe in “telecommunicative” contact with the IPG; and the clinicianprogrammer is likewise considered to be in telecommunicative contactwith the IPG, e.g., through the handheld programmer). The charging andforward telemetry circuitry 172 demodulates the carrier signal itreceives through the coil 170 to recover the programming data, e.g., theoperating program and/or the stimulation parameters, which programmingdata is then stored within the memory 162, or within other memoryelements (not shown) distributed throughout the IPG 100.

The microcontroller 160 is further coupled to monitoring circuits 174via bus 173. The monitoring circuits 174 monitor the status of variousnodes or other points 175 throughout the IPG 100, e.g., power supplyvoltages, current values, temperature, the impedance of electrodesattached to the various electrodes E1 . . . En, and the like.Informational data sensed through the monitoring circuit 174 may be sentto a remote location external to the IPG (e.g., a non-implantedlocation) through back telemetry circuitry 176, including a transmissioncoil 177.

The operating power for the IPG 100 may be derived from a rechargeablepower source 180 according to an exemplary embodiment of the presentinvention. The rechargeable power source 180 may comprise a lithium-ionor lithium-ion polymer battery, for example. The rechargeable battery180 provides an unregulated voltage to power circuits 182. The powercircuits 182, in turn, generate the various voltages 184, some of whichare regulated and some of which are not, as needed by the variouscircuits located within the IPG 100.

In one exemplary embodiment, any of the n electrodes (En) may beassigned to up to k possible groups (where k is an integer correspondingto the number of channels). In one preferred embodiment, k may be equalto 4. Moreover, any of the n electrodes can operate, or be included in,any of the k channels. The channel identifies which electrodes areselected to synchronously source or sink current to create an electricfield. Amplitudes and polarities of electrodes on a channel may vary,e.g., as controlled by the patient hand held programmer 202. Externalprogramming software in the clinician programmer 204 is typically usedto set parameters including electrode polarity, amplitude, pulse rateand pulse width for the electrodes of a given channel, among otherpossible programmable features.

Hence, it is seen that each of the n programmable electrode contacts canbe programmed to have a positive (sourcing current), negative (sinkingcurrent), or off (no current) polarity in any of the k channels.Moreover, it is seen that each of the n electrode contacts can operatein a bipolar mode or multipolar mode, e.g., where two or more electrodecontacts are grouped to source/sink current at the same time.Alternatively, each of the n electrode contacts can operate in amonopolar mode where, e.g., the electrode contacts associated with achannel are configured as cathodes (negative), and the case electrode,on the IPG case, is configured as an anode (positive).

Further, according to an exemplary embodiment, the amplitude of thecurrent pulse being sourced or sunk from a given electrode contact maybe programmed to one of several discrete current levels. For example,the current pulse may be programmed to one of several discrete currentlevels between 0 to 10 mA, in steps of 0.1 mA. Also, in the exemplaryembodiment, the pulse width of the current pulses is adjustable inconvenient increments. For example, the pulse width range may be 0 to 1milliseconds (ms) in increments of 10 microseconds (μs). Similarly, inthe preferred embodiment, the pulse rate is adjustable within acceptablelimits. For example, the pulse rate preferably spans 0-1000 Hz. Otherprogrammable features can include slow start/end ramping, burststimulation cycling (on for X time, off for Y time), and open or closedloop sensing modes.

The stimulation pulses generated by the IPG 100 may be charged balanced.This means that the amount of positive charge associated with a givenstimulus pulse are offset with an equal and opposite negative charge.Charge balance may be achieved through a coupling capacitor, whichprovides a passive capacitor discharge that achieves the desired chargebalanced condition. Alternatively, active biphasic or multi-phasicpulses with positive and negative phases that are balanced may be usedto achieve the needed charge balanced condition.

Advantageously, the IPG 100 is able to individually control the nelectrode contacts associated with the n electrode nodes E1, E2, E3, . .. En. Controlling the current sources 186 and switching matrix 188 usingthe microcontroller 160, in combination with the control logic 166 andtimer logic 168, thereby allows each electrode contact to be paired orgrouped with other electrode contacts, including the monopolar caseelectrode, to control the polarity, amplitude, rate, pulse width andchannel through which the current stimulus pulses are provided. Otheroutput circuits can be used with the invention, including voltageregulated output, multiplexed channels, and the like.

As shown in FIG. 3, much of circuitry included within the IPG 100 may berealized on a single application specific integrated circuit (ASIC) 190.This allows the overall size of the IPG 100 to be quite small, andreadily housed within a suitable hermetically-sealed case. The IPG 100may include n feedthroughs to allow electrical contact to beindividually made from inside of the hermetically-sealed case with the nelectrodes that form part of the lead system outside of the case.

As noted earlier, in use, the IPG 100 may be placed in a surgically-madepocket, e.g., in the abdomen or just at the top of the buttocks, anddetachably connected to the lead system (comprising optional leadextension 120 and electrode array 110). While the lead system isintended to be permanent, the IPG 100 may be replaced should its powersource fail, or for other reasons.

The back telemetry features of the IPG 100 allow the status of the IPGto be checked. For example, when the external hand-held programmer 202(and/or the clinician programmer 204), initiates a programming sessionwith the implant system 10 (FIG. 1), the capacity of the battery istelemetered so that the external programmer can calculate the estimatedtime to recharge. Any changes made to the current stimulus parametersare confirmed through back telemetry, thereby assuring that such changeshave been correctly received and implemented within the implant system.Moreover, upon interrogation by the external programmer, allprogrammable settings stored within the implant system 10 may beuploaded to one or more external programmers.

Turning next to FIG. 4, a hybrid block diagram of an alternativeembodiment of an IPG 100′ that may be used with the invention isillustrated. The IPG 100′ includes both analog and digital dies, orintegrated circuits (ICs), which may be housed in a singlehermetically-sealed rounded case having, for instance, a diameter ofabout 45 mm and a maximum thickness of about 10 mm. Many of the circuitscontained within the IPG 100′ are identical or similar to the circuitscontained within the IPG 100, shown in FIG. 3. The IPG 100′ includes aprocessor die, or chip, 160′, an RF telemetry circuit 172′ (typicallyrealized with discrete components), a charger coil 170′, a lithium ionor lithium ion polymer battery 180′, battery charger and protectioncircuits 182′, memory circuits 162′ (SEEPROM) and 163′ (SRAM), a digitalIC 191′, an analog IC 190′, and a capacitor array and header connector192′.

The capacitor array and header connector 192′ include sixteen outputdecoupling capacitors, as well as respective feed-through connectors forconnecting one side of each decoupling capacitor through thehermetically-sealed case to a connector to which the electrode array110, or lead extension 120, may be detachably connected.

The processor 160′ may be realized with an application specificintegrated circuit (ASIC), field programmable gate array (FPGA), or thelike that comprises a main device for full bi-directional communicationand programming. The processor 160′ may utilize an 8086 core (the 8086is a commercially-available microprocessor available from, e.g., Intel),or a low power equivalent thereof, 16 kilobytes of SRAM memory, twosynchronous serial interface circuits, a serial EEPROM interface, and aROM boot loader 735. The processor die 160′ may further include anefficient clock oscillator circuit 164′ and a mixer andmodulator/demodulator circuit implementing the QFAST RF telemetry methodsupporting bi-directional telemetry at 8 Kbits/second. QFAST stands for“Quadrature Fast Acquisition Spread Spectrum Technique,” and representsa known and viable approach for modulating and demodulating data. Ananalog-to-digital converter (A/D) circuit 734 is also resident on theprocessor 160′ to allow monitoring of various system level analogsignals, impedances, regulator status and battery voltage. The processor160′ further includes the necessary communication links to otherindividual ASICs utilized within the IPG 100′. The processor 160′, likeall similar processors, operates in accordance with a program that isstored within its memory circuits.

The analog IC (AIC) 190′ may comprise an ASIC that functions as the mainintegrated circuit that performs several tasks necessary for thefunctionality of the IPG 100′, including providing power regulation,stimulus output, and impedance measurement and monitoring. Electroniccircuitry 194′ performs the impedance measurement and monitoringfunction.

The analog IC 190′ may also include a number of output current sources186′. The output current sources 186′ are configured to supply currentto a load, such as tissue, for example. The output current sources 186′may be configured to deliver up to 20 mA aggregate and up to 12.7 mA ona single channel in 0.1 mA steps. However, it will be noted that theoutput current sources 186′ may be configured to deliver any amount ofaggregate current and any amount of current on a single channel,according to one exemplary embodiment. The output current sources 186′will be described in more detail below.

Regulators for the IPG 100′ supply the processor and the digitalsequencer with a voltage. Digital interface circuits residing on theanalog IC 190′ are similarly supplied with a voltage. A programmableregulator supplies the operating voltage for the output current sources186′, e.g., sixteen bi-directional output current sources, each of whichmay be configured to either source or sink current as previouslymentioned. Each output current source 186′ may be connected to anelectrode node (En; FIG. 3). Each electrode node, in turn, may beconnected to a coupling capacitor Cn (FIG. 3). The coupling capacitorsCn and electrode nodes, as well as the remaining circuitry on the analogIC 186′, may all be housed within the hermetically sealed case of theIPG 100. A feedthrough pin, which is included as part of the headerconnector 192′, allows electrical connection to be made between each ofthe coupling capacitors Cn and the respective electrodes E1, E2, E3, . .. , or E16, to which each output current source 186′ is associated.

The digital IC (DigIC) 191′ functions as the primary interface betweenthe processor 160′ and the AIC output circuits 186′. The main functionof the DigIC 191′ is to provide stimulus information to the outputcurrent sources 186′. The DigIC 191′ thus controls and changes thestimulus levels and sequences when prompted by the processor 160′. In anexemplary embodiment, the DigIC 191′ comprises a digital applicationspecific integrated circuit (digital ASIC).

In one exemplary embodiment of the present invention, each of the outputcurrent sources 186′ of FIG. 4 may include a current digital to analogconverter (DAC) configured to regulate the current that is delivered tothe load. As mentioned previously, the current DAC is located in serieswith the load. FIG. 5 illustrates an exemplary output current source 500and a corresponding output current sink 501 each having current DACcircuitry 502, 503 in series with load 505. The load 505 may be tissueor any other resistive load to which current is supplied.

As shown in FIG. 5, the output current source 500 may include a currentgenerator 506 configured to generate a reference current, I_(ref). Asuitable current generator is disclosed in U.S. Pat. No. 6,181,969 ('969patent), issued Jan. 30, 2001 in the name of inventor John C. Gord,which is incorporated herein by reference in its entirety. The currentI_(ref) is input into DAC circuitry 502, 503 which is configured toregulate and/or amplify I_(ref) and to output an output current I_(out).The relation between I_(out) and I_(ref) is determined in accordancewith input bits arriving on bus 513, which gives DAC circuitry 502 itsdigital-to-analog functionality. Essentially, in accordance with thevalues of the various M bits on bus 513, any number of output stages 514(i.e., M1) are tied together in parallel such that I_(out) can rangefrom I_(ref) to 2^(M)*I_(ref). (Fractional values of I_(ref) are alsopossible, as disclosed in the '969 patent, but such subtlety is ignoredherein for simplicity). Although not shown in FIG. 5, the output stages514 can contain other components, such as choke transistors and othertransistors designed to ensure good current matching in the currentmirror circuitry. However, as such structures are explained in theabove-incorporated '969 patent, they are not discussed further, and thesimpler view of DAC circuitry of FIG. 5 is used as a basis fordiscussion.

As noted above, the circuit of FIG. 5 also comprises an output currentsink 501 corresponding to the output current source 500. The outputcurrent sink 501 in FIG. 5 preferably includes DAC circuitry 503, whichis essentially similar in design to the DAC circuitry 502 of the outputcurrent source 500, although preferably formed of N-type MOS transistorsM2, M4.

The output current source 500 is coupled to an electrode 504 a, whilethe output current sink 501 is coupled to a different electrode 504 b.As explained in the above-incorporated '969 patent, an electrode willtypically be hard-wired to both an output current source 500 and anoutput current sink 501, only one (or neither) of which is activated ata particular time to allow the electrode to selectively be used aseither a source or sink (or as neither). The source and sink hard-wiredat each electrode are sometimes respectively referred to as PDACs andNDACs, reflecting the fact that the sources are typically formed ofP-type transistors while the sinks are typically formed of N-typetransistors. (The use of transistors of these polarities is sensiblegiven that the source is biased to a high voltage (V+) on the analog IC190′, where P-type transistors are most logical, while the sink isbiased to a low voltage (V−), where N-type transistors are most logical.The substrate connection (not shown) for the transistors would typicallybe tied to the appropriate power supply, either V+ or V−, but could alsobe tied to the source). Thus, as shown in FIG. 5, output current source500 may be associated with electrode 5 (i.e., 504 a) at a particularpoint in time, while output current sink 501 may be associated withelectrode 7 (i.e., 504 b) at that time. At a later time, electrodes 5and 7 could be switched such that 5 now operates as the sink, while 7operates as the source.

The M1/M3 and M2/M4 current mirrors require that transistors M1 and M2operate in a saturation mode, such that the channels of the transistorsare in “pinch off.” When in a saturation mode, the output currentI_(out) is proportional to the gate voltage of the transistors M1 or M2,but does not depend upon the drain voltage to the first order. However,to keep the transistors M1 and M2 in the saturation mode, a certaindrain-to-source voltage, V_(DS), has to be satisfied for eachtransistor. Specifically, V_(DS) must be greater than the gate-to-sourcevoltage (V_(GS)) minus the threshold voltage (V_(T)) of the transistor(i.e., V_(DS)>V_(GS)−V_(T)). (This saturation condition is necessarilysatisfied because V_(DS)=V_(GS) by virtue of the common gate/drainconnection of transistors M3 and M4). The minimum drain-to-sourcevoltage V_(DS) that satisfies this relationship and allows transistorsM1 and M2 to operate in the saturation mode is typically 0.5 to 0.7volts.

Assuming the load 505 has a resistance of R, the voltage drop across theload 505, V_(L), is equal to I_(out)*R. Because the transistors M1 andM2 are in series with the load 505, the total voltage drop, V_(TOT), dueto the output current source 500, the output current sink 501, and theload 505 may represented by the following equation:(V+)−(V−)=V_(TOT)=V_(DS1)+V_(L)+V_(DS2). The total power, P_(TOT),consumed by the circuitry shown in FIG. 5 is equal to V_(TOT)*I_(out).(It will be recognized by one skilled in the art that additional powermay be consumed by transistors M3 and M4 and other circuitry that may beincluded in the output current source 500 and sink 501. However, forpurposes of explaining the present invention, the power consumed by thecircuitry shown in FIG. 5 will be represented by the equationP_(TOT)=V_(TOT)*I_(out)). In other words, the total power consumed bythe DACs 502, 503 (and ultimately by the IPG 100) is in part determinedby the voltage drops V_(DSx) that occur across transistors M1 and M2,which as just noted are significant. Accordingly, the life of thebattery or other power source for the IPG 100 is adversely affected bythese voltage drops. Hence, a current DAC design that results in areduction in the drain-to-source voltage drops V_(DS1) and V_(DS2) of M1and M2 is highly desirable to reduce power consumption and prolongbattery life.

FIG. 6 illustrates an embodiment of the present invention in which anoutput current source 600 and its corresponding output current sink 610respectively comprise DAC circuitry 601, 611 configured to regulateI_(ref) and deliver a regulated and/or amplified current I_(out) to loadR. The DAC circuitry 601, 611 includes MOS transistors M1, M2 that donot have to operate in a saturation mode to provide the regulated outputcurrent I_(out). According to one exemplary embodiment, the outputcurrent source/sink 600, 610 may be used as the output currentsource/sink 186′ in the IPG 100′ of FIG. 4.

While both the output current source and sink 600, 610 are shown in FIG.6, it should be understood that the invention can comprise either ofthese circuits separately. Moreover, output current source 600 could beused with a different output current sink than that shown as 610, orcould be used with no corresponding output current sink. Likewise,output current sink 610 could be used with a different output currentsource than that shown as 600, or could be used with no correspondingoutput current source. However, because it is preferred to use both asource and sink together in an IPG application, and because it ispreferred for convenience that the designs be symmetrical for each, thesource and sink are so depicted in FIG. 6.

As shown in FIG. 6, the DAC circuitry 601, 611 in one embodiment issimilar in design, and preferably comprises P-type MOS transistors inthe output current source 600 and N-type MOS transistors in the outputcurrent sink 610. However, this is not strictly required, and devices ofdifferent polarities could be used in either the source 600 or sink 610with the same advantages. As with the output current source/sink 500/510of FIG. 5, the output stages 614 can be multiplied in conjunction withthe status of the digital bits on bus 613 such that I_(out) can rangefrom I_(ref) to 2^(M)*I_(ref). However, because the analog aspects ofthe DAC circuitry are most important to illustration of embodiments ofthe invention, such analog aspects are the focus of the circuitrydepicted in FIG. 6. The digital aspects of the DAC circuitry arediscussed in further detail in the above-incorporated '969 patent.

DAC circuitry 601 or 611 in the embodiment of FIG. 6 comprises a MOStransistor (M1 or M2), an operational amplifier (op amp) (A1 and A2),and two resistors (R₁ and R₂, or R₃ and R₄). Because believed easiest toexplain, the operation of the output current sink 610 is describedbelow. It should be understood that the output current source 600 wouldoperate using similar principles, although modified to account forchanges in the polarity of the circuits used compared to the sink 610.Moreover, while the use of an N-type MOS transistor M2 is shown for usein the output current sink 610, it is again worth mentioning thatdevices of different polarities can be used in other useful embodiments.

As shown in FIG. 6, a pair of resistors R₁ and R₂ matched with aresistive ratio N are coupled together through a negative feedbacknetwork. Because the operational amplifier A2 has a very high impedance,V_(REF) (i.e., I_(ref)*R₁) is propagated to V_(r) by the feedbacknetwork: should V_(r) fall below V_(REF), then the output of op amp A2would be increased, which would tend to turn transistor M2 further on.This increases I_(out), which in turn increases V_(r) (i.e.,I_(out)*R₂), at least until such time as V_(r) would exceed V_(REF) atwhich point the feedback loop would have the opposite effect. Because ina steady state V_(REF)=V_(r), it can be seen that the ratio N between R₁and R₂ defines the current gain of the DAC circuitry 610 (i.e.,I_(out)=N*I_(ref)).

Importantly, transistor M2 does not have to operate in saturation modeto provide the regulated output current, I_(out). Thus, thedrain-to-source voltage drop V_(DS2) of M2 may be significantly lowerthan the drain-to-source voltage drop V_(DS2) of M2 in FIG. 5—less than0.2 volts, for example. The same is true for transistor M1 in thecurrent output source 600, which will experience a drop V_(DS2) of lessthan 0.2 volts, again less than corresponding transistor M1 in FIG. 5.While resistors R3 and R2 are also in the output stage, and thereforewill result in voltage drops that consume some power, such voltage dropswill normally be minimal because these resistances will generally by atleast an order of magnitude smaller than the “on” resistances of thetransistors M1 and M2 and the load R. For example, R3 and R2 may be ohmsor tens of ohms, which would be negligible compared to the “on”resistances of transistors M1 and M2, as well as load R (typically onthe order of hundreds or thousands of ohms). Thus, the DAC circuitrytopology of FIG. 6 advantageously reduces the amount of power that isconsumed by the output current source 600. It is important to note that,according to a preferred embodiment, the drain-to-source voltage dropV_(DS) may be any voltage that is less than the drain-to-source voltagedrop required for transistors M1, M2 to operate in saturation mode.

An additional advantage of the current DAC circuitry topology of FIG. 6is that the amplification of the input current I_(ref) may be easilyadjusted by adjusting the values of R₁ and R₂ (and R₃ and R4), which arerelated by the ratio, N. As mentioned previously, the output current,I_(out), is equal to N times the input current, I_(ref). Therefore, thecurrent gain may be increased merely by changing the ratio N between R₁and R₂ (and R₃ and R4). In contrast, multiple stages 514 of transistorsmust be added to the current DAC topology of FIG. 5 to amplify the inputcurrent, I_(ref). For example, FIGS. 7A and 7B respectively show theoutput current sources 500 and 600 of FIGS. 5 and 6 for exemplary outputcurrents, I_(out), of I_(ref), 2I_(ref), and 4I_(ref). In the design ofFIG. 5 (FIG. 7A), increasing the current requires multiplying inparallel the number of devices (i.e., transistors) in output stages 514by a factor of 2^(M). In other words, at least 2^(M) devices(potentially hundreds or thousands) must be fabricated on the analog IC190′. (See FIGS. 8A and 8B of the above-incorporated '969 patent andassociated text for further details). By contrast, when the design ofFIG. 6 is used (FIG. 7B), the number of devices in the output stages 614does not change as higher output currents are called for; instead, onlythe values of the resistors (R₁ and R₂; R₃ and R₄) change. Thus, thepreferred current DAC circuitry topology of FIG. 6 simplifies overallIPG 100 design and reduces layout requirements.

FIG. 8 illustrates a preferred alternative embodiment of the presentinvention in which an output current source 800 and an output currentsink 810 comprise an alternative DAC circuitry topology 801, 811configured to regulate I_(ref) and deliver a regulated and/or amplifiedcurrent I_(out) to a load, R. As with the embodiment of FIG. 6, anembodiment of the invention can comprise either circuit 800 or 810individually, although in an IPG application it is preferred to use twosimilarly-designed sources/sinks together. Moreover, the polarities ofthe devices used in both the output current source 800 and outputcurrent sink 810 can be changed from what is depicted. Because believedeasiest to explain, the operation of the output current sink 810 isdescribed below, although it should be understood that the symmetricoutput current source 800 would operate using similar principles,although modified to account for changes in the polarity of the devicesused.

As will be explained in detail below, the DAC circuitry 811 of theoutput current sink 810 preferably includes N-type MOS transistors M2,M5, and M6 that, like the embodiment of FIG. 6, do not have to operatein saturation mode to provide the regulated output current I_(out). MOStransistors M5 and M6 form matched active loads. Op Amp A2 andtransistor M2 are configured to act as a feedback loop to control thedrain voltage of M2, resulting in V_(r)=V_(REF). In a preferredembodiment, the common gate of M5 and M6 are biased (via V_(GS)) suchthat M5 and M6 operate in a linear mode as opposed to a saturation mode,and hence have small voltage drops from source to drain. Hence, thetotality of the drain-to-source drops (i.e., across M2 and M6) may besignificantly lower than the drain-to-source voltage drop V_(DS1) of M2in FIG. 5—less than 0.2 volts, for example. The same is true fortransistors M1, M4 in the output current source 800, which are likewisenot in saturation. Thus, the preferred current DAC circuitry topology801, 811 of FIG. 8 advantageously reduces the amount of power that isconsumed by the output current source 800 and/or the output current sink810.

In one embodiment, transistors M5 and M6 (and M3 and M4) are identical,and hence have identical drain-to-source resistances. However, in otheruseful embodiments, these drain-to-source resistances can be modified toachieve the benefits of FIG. 6, such as increased output current gainsusing simplified layouts, as discussed earlier. For example, assume thattransistor M5 is N times more resistive than transistor M6, for example,by increasing M5's channel length by N or decreasing M5's channel widthby N. The two transistors M5, M6 of FIG. 8 would then stand in the samerelationship as do resistors R₁ and R₂ of FIG. 6, thus allowing thecircuit to provide an output current gain that is N times larger thanthe reference current (I_(out)=N*I_(ref)).

As with the embodiment of FIG. 6, the embodiment of FIG. 8 may beextended to operate as a multi-bit DAC, although such structures (likebit bus 613, stages 614, etc.) are not shown in FIG. 8 for convenience.

The approaches of FIGS. 6 and 8 are generically illustrated in FIG. 9,which shows a generic structure for an output current source 900 and anoutput current sink 910. The DAC circuitry 901, 911 for each comprisestwo loads (Z₁ and Z₂; Z₃ and Z₄), and a feedback circuit, H forcontrolling the gate of the transistor M1 or M2 so as to operate thattransistor in a linear mode, but not a saturation mode. In fact, thecircuits can even be further generalized when it is realized that theloads Z_(x) can comprise part of the feedback circuit H. What isimportant is to receive feedback concerning the mode in which thetransistors M1 or M2 are operating so that these transistors can becontrolled to operate in a linear mode, which is most easilyaccomplished by monitoring the voltages at nodes on the input path 920and the output path 921. As several such circuits are possible ofachieving this result, it should be understood that the circuits ofFIGS. 6 and 8 are merely exemplary.

It should be understood that the direction in which current flows is arelative concept, and different conventions can be used to definewhether currents flow to or from various sources. In this regard, arrowsshowing the directions of current flows in the Figures, references tocurrent flowing to or form various circuit nodes, references to currentsbeing sunk or sourced, etc., should all be understood as relatively andnot in any limiting sense.

It should also be understood that reference to an “electrode”implantable adjacent to a tissue to be stimulated includes electrodes onthe implantable stimulator device, or associated electrode leads, or anyother structure for stimulating tissue.

While the invention herein disclosed has been described by means ofspecific embodiments and applications thereof, numerous modificationsand variations could be made thereto by those skilled in the art withoutdeparting from the literal and equivalent scope of the invention setforth in the claims.

1. An implantable stimulator device, comprising: a first electrodeimplantable adjacent to tissue to be stimulated; and a first stage in afirst output current source or sink coupled to the first electrode,wherein the first stage comprises a first transistor for passing anoutput current along a first output path to the first electrode, whereinthe drain-to-source voltage drop across the first transistor is lessthan a drain-to-source voltage drop of the first transistor whenoperating in a saturation mode, wherein the output current is set by areference current input to the first stage, and wherein the outputcurrent equals the reference current.
 2. The device of claim 1, furthercomprising: a second electrode implantable adjacent to the tissue to bestimulated; and a second stage in a second output current sink or sourcecomprising the other of the source or sink of the first output currentsource or sink, the second stage coupled to the second electrode,wherein the second output current sink or source comprises a secondtransistor for receiving the output current along a second output pathfrom the second electrode, wherein the drain-to-source voltage dropacross the second transistor is less than a drain-to-source voltage dropof the second transistor when operating in a saturation mode.
 3. Thedevice of claim 1, wherein the first stage further comprises anamplifier for driving a gate of the first transistor.
 4. The device ofclaim 3, wherein the inputs to the amplifier comprise a referencevoltage and a voltage fed back from the first output path.
 5. The deviceof claim 1, wherein a gate of the first transistor is influenced by avoltage fed back from the first output path.
 6. The device of claim 1,wherein the magnitude of the output current compared to the referencecurrent is set by digital bits sent to the first output current sourceor sink.
 7. At least one output current source or sink for animplantable stimulator device, comprising: a first input path forreceiving a reference current; a first output path for providing anoutput current to a first electrode of the implantable stimulatordevice, wherein the first electrode is implantable adjacent to tissue tobe stimulated, and wherein a magnitude of the output current is dictatedby a magnitude of the reference current; and a first feedback circuitfor comparing a first voltage on the first input path and a secondvoltage on the first output path, wherein the first feedback circuitcontrols a first transistor in the first output path such that the firsttransistor operates in a linear mode and not in a saturation mode. 8.The device of claim 7, further comprising another output current sourceor sink to complement the first output current source or sink,comprising: a second input path for receiving a reference current; asecond output path for receiving the output current to a secondelectrode of the implantable stimulator device, wherein the secondelectrode is implantable adjacent to the tissue to be stimulated, andwherein a magnitude of the output current is dictated by a magnitude ofthe reference current; a second feedback circuit for comparing a firstvoltage on the second input path and a second voltage on the secondoutput path, wherein the second feedback circuit controls a secondtransistor in the second output path such that the second transistoroperates in a linear mode and not in a saturation mode.
 9. The device ofclaim 7, wherein the output current equals the reference current. 10.The device of claim 7, wherein the output current is N times higher thanthe reference current.
 11. The device of claim 10, wherein N is providedby first and second loads whose impedances vary by the ratio N.
 12. Thedevice of claim 7, wherein the feedback circuit comprises an amplifierwhich outputs a control signal to a gate of the transistor.
 13. Thedevice of claim 7, wherein the first voltage is generated by passing thereference current through a first load, and wherein the second voltageis generated by passing the output current through a second load. 14.The device of claim 7, wherein the magnitude of the output currentcompared to the magnitude of the reference current is set by digitalbits sent to the first output current source or sink.
 15. At least oneoutput current source or sink for an implantable stimulator device,comprising: a first input path comprising a first load coupled to apower supply for receiving a reference current; a first output pathcomprising a second load coupled to the power supply and a transistor,wherein the first output path comprises an output current coupled to afirst electrode of the implantable stimulator device, wherein the firstelectrode is implantable adjacent to tissue to be stimulated; and afirst feedback circuit for comparing a first voltage on the first inputpath and a second voltage on the first output path, wherein the firstfeedback circuit controls the transistor such that the transistoroperates in a linear mode and not in a saturation mode.
 16. The deviceof claim 15, further comprising another output current source or sink tocomplement the first output current source or sink, comprising: a secondinput path comprising a first load coupled to a power supply forreceiving a reference current; a second output path comprising a secondload coupled to the power supply and a transistor, wherein the secondoutput path comprises an output current coupled to a second electrode ofthe implantable stimulator device, wherein the second electrode isimplantable adjacent to the tissue to be stimulated; and a secondfeedback circuit for comparing a first voltage on the second input pathand a second voltage on the second output path, wherein the secondfeedback circuit controls the transistor such that the transistoroperates in a linear mode and not in a saturation mode.
 17. The deviceof claim 15, wherein the output current equals the reference current.18. The device of claim 15, wherein the output current is N times higherthan the reference current.
 19. The device of claim 18, wherein N isprovided by the first and second loads whose impedances vary by theratio N.
 20. The device of claim 15, wherein the first and second loadsare active transistors with a common bias voltage.
 21. The device ofclaim 15, wherein the first and second loads are passive resistors. 22.The device of claim 15, wherein the feedback circuit comprises anamplifier which outputs a control signal to a gate of the transistor.23. The device of claim 15, wherein the magnitude of the output currentcompared to the magnitude of the reference current is set by digitalbits sent to the first output current source or sink.